PHOTODIODE PREAMP

The performance of the photodiode preamp shown in Figure 7

is enhanced by the AD548’s low input current, input voltage

offset, and offset voltage drift. The photodiode sources a current

proportional to the incident light power on its surface. RF converts

the photodiode current to an output voltage equal to RF × IS.

Application Hints–AD548

Figure 7.

An error budget illustrating the importance of low amplifier

input current, voltage offset, and offset voltage drift to minimize

output voltage errors can be developed by considering the equi-

valent circuit for the small (0.2 mm2 area) photodiode shown in

Figure 7. The input current results in an error proportional to

the feedback resistance used. The amplifier’s offset will produce

an error proportional to the preamp’s noise gain (I + RF/RSH),

where RSH is the photodiode shunt resistance. The amplifier’s

input current will double with every 10°C rise in temperature,

and the photodiode’s shunt resistance halves with every 10°C

rise. The error budget in Figure 8 assumes a room temperature

photodiode RSH of 500 MΩ, and the maximum input current

and input offset voltage specs of an AD548C.

TEMP

؇C RSH (M⍀)

VOS (V) (1+ RF/RSH) VOS IB (pA) IBRF

TOTAL

–25 15,970

0 2,830

25 500

50 88.5

75 15.6

85 7.8

150

200

250

300

350

370

151 µV

207 µV

300 µV

640 µV

2.6 mV

5.1 mV

0.30

2.26

10.00

56.6

320

640

30 µV 181 µV

262 µV 469 µV

1.0 mV 1.30 mV

5.6 mV 6.24 mV

32 mV 34.6 mV

64 mV 69.1 mV

Figure 8. Photodiode Preamp Errors Over Temperature

The capacitance at the amplifier’s negative input (the sum of the

photodiode’s shunt capacitance, the op amp’s differential input

capacitance, stray capacitance due to wiring, etc.) will cause a

rise in the preamp’s noise gain over frequency. This can result in

excess noise over the bandwidth of interest. CF reduces the

noise gain “peaking” at the expense of bandwidth.

INSTRUMENTATION AMPLIFIER

The AD548C’s maximum input current of 10 pA makes it an

excellent building block for the high input impedance instru-

mentation amplifier shown in Figure 9. Total current drain for

this circuit is under 600 µA. This configuration is optimal for

conditioning differential voltages from high impedance sources.

The overall gain of the circuit is controlled by RG, resulting in

the following transfer function:

VOUT = 1 + (R1 + R2 )

VIN RG

Figure 9. Low Power Instrumentation Amplifier

Gains of 1 to 100 can be accommodated with gain nonlinearities

of less than 0.01%. Input errors, which contribute an output

error proportional to in amp gain, include a maximum untrimmed

input offset voltage of 0.5 mV and an input offset voltage drift

over temperature of 4 µV/°C. Output errors, which are indepen-

dent of gain, will contribute an additional 0.5 mV offset and

4 µV/°C drift. The maximum input current is 15 pA over the

common-mode range, with a common-mode impedance of over

1 × 1012 Ω. Resistor pairs R3/R5 and R4/R6 should be ratio

matched to 0.01% to take full advantage of the AD548’s high

common-mode rejection. Capacitors C1 and C1′ compensate for

peaking in the gain over frequency caused by input capacitance

when gains of 1 to 3 are used.

The –3 dB small signal bandwidth for this low power instrumenta-

tion amplifier is 700 kHz for a gain of 1 and 10 kHz for a gain of

100. The typical output slew rate is 1.8 V/µs.

LOG RATIO AMPLIFIER

Log ratio amplifiers are useful for a variety of signal conditioning

applications, such as linearizing exponential transducer outputs

and compressing analog signals having a wide dynamic range.

The AD548’s picoamp level input current and low input offset

voltage make it a good choice for the front-end amplifier of the

log ratio circuit shown in Figure 10. This circuit produces an

output voltage equal to the log base 10 of the ratio of the input

currents I1 and I2. Resistive inputs R1 and R2 are provided for

voltage inputs.

Input currents I1 and I2 set the collector currents of Q1 and Q2,

a matched pair of logging transistors. Voltages at points A and

B are developed according to the following familiar diode

equation:

VBE = (kT/q) ln (IC /IES )

In this equation, k is Boltzmann’s constant, T is absolute tem-

perature, q is an electron charge, and IES is the reverse saturation

current of the logging transistors. The difference of these two

voltages is taken by the subtractor section and scaled by a factor

of approximately 16 by resistors R9, R10, and R8. Temperature

REV. D

–9–